1. Field of the Invention
The present invention relates generally to closed loop power control systems, and relates more particularly to closed loop power control systems with reduced EMI and switching losses and control of the rate of change of gate drive signals.
2. Description of Related Art
High-speed switching devices such as bipolar transistors, MOSFETs and IGBT's enable increased carrier frequency for voltage-source PWM inverters, thus leading to much better operating characteristics. High-speed switching, however, causes the following serious problems, originating from a high rate-of-change in voltage and/or current:
a) ground current escaping to earth through stray capacitors inside motors and through long cables;
b) conducted and radiated EMI;
c) motor bearing current and shaft voltage; and
d) shortening of insulation life of motors and transformers.
The voltage and/or current change caused by high-speed switching produces high-frequency oscillatory common-mode and differential-mode currents when the switching device(s) change state because parasitic stray capacitance inevitably exists inside a load, for example, an ac motor, as well as inside the switching converter. Thus, each time an inverter switching event occurs, the potential of the corresponding inverter output terminal moves rapidly with respect to ground, and a pulse of common mode current flows in the d-c link to the inverter, via the capacitance of the heatsink motor cable and motor windings to ground. The amplitude of this pulse of current for a class B (residential) motor drive is typically several hundred millamps to several amps; and the pulse width is typically 250 to 500 ns. For a class A drive (Industrial), and depending on the size of the motor and length of the motor cable, the pulse current amplitude is typically several amperes with a pulse width of 250 ns to 500 ns, to 20 amperes or more with a pulse width of 1 to 2 μs.
The common mode oscillatory currents may have a frequency spectrum range from the switching frequency of the converter to several tens of MHZ, which creates a magnetic field and will produce radiated electromagnetic interference (EMI) throughout, thus adversely affecting electronic devices such as radio receivers, medical equipment, etc.
A number of Governmental restrictions apply to the degree of permissible line current EMI and permissible ground current in certain motor applications. Thus, in Class B residential (appliances), applications, ground current must be kept below from 1 to 20 mA over a frequency range from 0 to 30 kHz respectively (over a logarithmic curve); and conducted line current EMI must be kept below designated values (less than about 60 dB*V) over a frequency range of 150 kHz to 300 MHZ. For motor drive applications designated as class A Industrial applications, limitations on ground current are less stringent, but line current EMI is still limited over the 150 kHz to 30 MHZ range.
Generally, common-mode chokes and EMI filters, based on passive elements, may not completely solve these noise and EMI problems. Passive filters, consisting of a common mode inductor and “Y” capacitors in the input ac line have been used to filter the common mode current in such motor drive circuits. Passive common mode filters may place limits on the PWM frequency which can be used, are physically large (frequently a major fraction of the volume of the motor drive structure) and are expensive. Further, they are functionally imperfect in that they exhibit undesired resonance which runs counter to the desired filtering action. Further, in general purpose industrial drives, the drive circuit and motor are often connected by cables which are up to 100 meters or more long. The longer the cable, the greater the conducted common mode EMI in the motor cable, and the larger the required size of a conventional passive common mode input filter.
A common-mode transformer with an additional winding shorted by a resistor is known which can damp the oscillatory ground current. Unfortunately, a small amount of aperiodic ground current will still remain in this circuit.
Active filters for control of the common mode current in a pulse width modulated (PWM) controlled motor drive circuit are well known. Typical devices are described in the paper an Active Circuit for Cancellation of Common-Mode Voltage Generated by a PWM Inverter, by Satoshi Ogasawara et al., IEES Transactions on Power Electronics, Vol. 13, No. 5, (September 1998 and in U.S. Pat. No. 5,831,842 in the names of Ogasawara et al.
FIG. 10 shows a typical prior art active filter circuit or EMI and noise cancellation device for an a-c motor. Thus, in FIG. 10, an a-c source comprising an input terminal L and a neutral terminal are connected to the a-c input terminals of a full wave bridge connected rectifier 40. While a single phase supply is shown, the principles in this and in all Figures to be described can be carried out with a three-phase or multi-phase input. The positive and negative busses of rectifier 40 contain points A and D respectively and are connected to a three-phase bridge connected PWM controlled inverter 41, at inverter terminals B and F. The output a-c terminals of the inverter are connected to a-c motor 42. A filter capacitor 40a is also connected across terminals B and F. Motor 42 has a grounded housing connected to ground wire 43 with ground terminal 43a. 
The active filter consists of a pair of transistors Q1 and Q2, connected across the d-c output lines of rectifier 40 with their emitters connected at node E. These define amplifiers which are controlled by output winding 44 of a differential transformer having input windings 45 and 46 connected in the positive and negative output busses of rectifier 40. The winding polarities are designated by the conventional dot symbols. Winding 44 is connected between the control terminals of transistors Q1 and Q2 and the common emitter node E. A d-c isolating capacitor 47 is connected to ground line 43 at node C.
The active filter including capacitor 47 defines a path for diverting the majority of the common mode current which can otherwise flow in the path L or N, A, B, M (motor 42), 43, 43a and back to L or N; (or in the reverse path when polarity reverses) or in path L or N, D, F, M, 43, 43a (or in the reverse path when polarity reverses). Thus, most common mode current can be diverted, for currents from positive terminal A, in the path B, M, C, E, Q2, F, B, for “positive current”, and in the pattern B, M, C, E, Q1, B for “negative” current. by the proper control of transistor Q1 and Q2. The path for common mode current flowing into negative terminal D follows the path F, M, C, E, Q2, F for “positive” current and F, M, C, E, Q1, B for “negative” current. The degree of diversion depends on the current gain of winding 44 and the current gain of Q2, for “positive current”, and the current gain of winding 44 and current gain of Q1, for “negative” current. In order to obtain a sufficient degree of diversion of the common mode current, the overall current gain of winding 44 and transistors Q1 and Q2 must be high.
The sensing transformer 44, 45, 46 of FIG. 10 has been large and expensive in order to provide sufficiently high current gain. It would be very desirable to reduce the size and cost of this transformer without jeopardizing the operation of the circuit. A further problem is that because of the high gain required, this closed-loop circuit has a tendency to produce unwanted oscillation.
Further, it has been found that the transistors Q1 and Q2 may not be able to operate in their linear regions over a large enough range within the “headroom” defined by the circuit, thus defeating the active filtering action. The headroom, or the voltage between the collector and emitter of transistors Q1 and Q2 is best understood by considering the approximate equivalent circuit of FIG. 10, as shown in FIG. 11, in which the ground potential at C is the same as that of the neutral line in FIG. 10. Transistors Q1 and Q2 are shown as resistors R1 and R2 respectively with respective parallel connected diodes. The d-c bridge 40 is shown as two d-c sources 50 and 51, each producing an output voltage of VDC/2 where VDC is the full output voltage between the positive and negative busses at terminals A and D, and an a-c source 52 having a peak a-c voltage of VDC/2.
It can be seen from FIG. 11 that headroom can disappear at different portions of the cycle of source 52. Thus, consider a first situation in which the leakage impedances of transistors Q1 and Q2 are the same. In this case, the values of resistors R1 and R2 in FIG. 2 are about equal. Now, as the ground potential at terminal C swings between (+)VDC/2 and (−)VDC/2 with respect to the d-c midpoint at node 53 in FIG. 2, the potential at the emitters of transistors Q1 and Q2 also swings between (+)VDC/2 and (−)VDC/2, if it is assumed that the impedance of capacitor 47 is much smaller than R1 and R2. Therefore, during the periods when the potential at node E is close or equal to the potential of the d-c bus (at points B or F), insufficient voltage headroom exists for the relevant transistors Q1 or Q2 to operate as linear amplifiers, and the active filtering action is lost.
The above described filters are well known in a number of electromagnetic applications, particularly in power transfer systems. Systems involving power transfer typically include power inverters that can be used for power supply applications in addition to motor drives. Power inverters are typically supplied with electrical power through power transmission lines that are operated in a multi-phase mode. For example, a three phase power supply is typical in applications involving inverter operation and motor drives. A three phase power supply includes three transmission lines with a voltage potential between the three pairs of power delivery lines. That is, if the three phase input is supplied through lines L1, L2 and L3, there is a voltage potential between lines L1 and L2, between lines L2 and L3, and between lines L1 and L3. These phase-to-phase voltages are typically sinusoidal and out of phase with respect to each other to provide efficient power transfer.
In a three phase system like that described above, the transmission lines act as differential voltage pairs in transmitting a power signal that is the value of the voltage between the various line pairs. This type of power transmission scheme is very useful in transmitting a power signal with immunity to noise interruptions that affect all the power lines at the same time. That is, if all of the power lines are impacted by a common interference or noise signal, all lines are affected to the same degree and the differential voltages remain the same. Accordingly, it is often the case that three phase transmission lines carry a common mode voltage that does not necessarily impact the power signals delivered to an inverter, for example.
When an inverter is used to power and control a motor drive system, the inverter typically uses high frequency switching to direct the appropriate power signals to the motor windings to produce the desired operation performance. For example, the inverter can be operated to control the motor for a specified torque operation, or a desired velocity. Due to the high frequency switching of the inverter, it is often the case that there are abrupt voltage transitions on the lines driving the motor, which are an inherent source of EMI. This EMI can produce common mode noise that causes interference in motor control signals, feedback signals I/O, sensors and the like. In addition, capacitive coupling with inverter outputs and ground, or the motor grounding by itself can produce high frequency ground currents that provide further interference with control signals and other communication signals. High frequency ground currents can also lead to radiated interference and produce ground loops that act as loop antennas to increase the production of radiated noise. The high frequency ground currents can also result in instantaneous voltage differences between two ground potential points, which interferes with appropriate references for control and communication signals.
In addition to the above mentioned filters, a number of measures are available to reduce and control common mode noise and radiated EMI. For example, shielded power cables are used to connect the inverter to the motor to prevent noise current from flowing out of the motor drive system to ground. The power lines to the motor are also twisted to provide a balanced capacitive coupling to reduce the stray capacitive coupling to ground. A common mode choke is often used on the power lines in the motor to attenuate the common mode noise as well. An EMI filter like that described above is often attached to the input of the inverter to act as a low pass filter to remove common mode noise from the earth ground that might otherwise create a ground voltage differential for one or more components of the motor drive system.
Another technique to reduce EMI noise is to measure high frequency noise current and provide compensation for any detected currents. As described above and in other prior art a current transformer has been used to sense noise current to determine appropriate compensation to control EMI. However, an appropriately sized and rated current transformer is bulky and expensive, and produces non-linear operation in practice. It would be desirable to provide a circuit and technique for reducing EMI without the use of a current transformer.
Often, an EMI reduction system is part of a large closed loop control for operating a synchronous motor with the inverter. For example, multiple high level systems can provide command and control signals to the inverter controller to operate the motor or power supply in conjunction with related high level systems. Accordingly, it would be desirable to reduce the EMI production of the overall system, in addition to the closed loop control involving the inverter and sensor feedback.
In the high voltage inverter system, level shifters are often used to provide control signals to the half bridges that make up the various stages of the motor drive inverter. In the level shifting system, references are changed typically from a logic voltage level to a reference level consistent with the inverter power supply. As a result, control signals are transmitted by the level shifting circuit in the form of pulses to avoid the additional energy losses resulting from power switches being maintained in a power conduction mode to permit signal transmission. Accordingly, an input signal is provided to a pulse generator that supplies a pulse train with a duty cycle representative of the input signal. The pulse train is then converted to a control signal for controlling the gate drive in the application. Often, due to the nature of high frequency, high power switching, voltage spikes on both the gate drive and half bridge components with a tremendous voltage change per unit time are observed. It would be desirable to reduce or control the change in voltage per unit time to prevent voltage spikes, which can result in excessive EMI and other disruptive control operations. Presently known solutions for controlling the dv/dt of high frequency high power switching applications are bulky, complex and costly. Accordingly, it would be desirable to obtain a simple control for high voltage, high frequency switching applications to modulate the dv/dt of inverter gate drives.
In a motor drive with an inverter, space vector modulation is often used to control the motor based on quadrant switching of switches in the inverter. In this type of motor control, accurate motor phase current measurements are useful to provide high performance control for a particular application, such as velocity or torque control. However, it is often difficult to accurately measure motor phase current over a wide current and temperature ranges. For example, Hall effect sensors can be used in the lines driving the motor, but they are inherently bulky and costly. In a pulse width modulated (PWM) inverter drive system, motor phase current can be determined from measurement of the d-c bus current when non zero basic vectors are used in the space vector modulation. Each basic vector is assigned a specific time in a PWM cycle to generate the command voltage vector. However, if a basic voltage vector is used only for a very short period of time, motor phase current cannot be directly determined from the d-c bus current. This lack of observability of motor phase current is due to practical considerations and limitations in the responsiveness of components of the PWM inverter drive system. For example, time delays caused by a/d converter sample and hold times, slewing of voltage during turn on, and other delay factors prevent the effects of basic vectors used for a very short time from being observed. Accordingly, it would be desirable to observe the effects of basic vectors used for a very short time, and to obtain overall values of motor phase currents for all control periods to achieve a high performance motor drive.